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Abstract
In this work, a new and efficient terahertz reflective phase shifter is proposed. The phase shifter is composed of a metal-dielectric-metal structure with a double dipole patch array, as well as copper grating electrodes immersed within the nematic liquid crystal. More specifically, the employed copper grating electrodes consist of two sets of cross-distributed comb grids, whereas at each set of comb grids can be applied an external bias voltage separately. On top of that, the electric field in the liquid crystal (LC) layer can be continuously changed by enforcing an innovative technique. Consequently, the orientation of the LC molecules was fully controlled by the applied electric field, since the dielectric constant of the LC is controlled by the biased voltage. The phase of the reflective electromagnetic wave can be continuously manipulated. Under this direction, the experimental results show that the phase shift exceeds the value of 180° in the range of 102.5 GHz-104.3 GHz, where the maximum phase shift is 249° at 103 GHz. The proposed work provides a new regulation concept for the implementation of LC-based terahertz devices and the respective applications in the terahertz reconfigurable antennas field.
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1. Introduction
In recent years, the rapid development of the terahertz (THz) technology in biomedicine [1,2], non-destructive testing [3,4], wireless communication [5,6], imaging [7–9], as well as other fields has attracted widespread attention. Along these lines, THz technology is regarded as an extremely important technology, since the development and utilization of its electromagnetic band is of both great scientific significance and potential application value [10]. Additionally, Liquid crystals (LCs) are widely employed for terahertz wave modulation due to their unique electromagnetic properties [11]. More specifically, LC is considered a tunable material with an extensive range of relative permittivity tunability properties from the millimeter-wave band to the THz frequency band [12,13]. As a result, a number of tunable devices have been realized under the application of an applied electric [14] or magnetic field [15]. These devices include phase shifters [16,17], modulators [18,19], absorbers [20,21], filters [22,23] and polarizers [24,25], etc. Reflective phase shifters are one of the key components for the fabrication of future terahertz-shaped reflector antennas and radio frequency (RF) devices [26,27].
Moreover, the conventional LC phase shifter adjusts the dielectric constant of the LC by loading an external voltage, which induces the LC permittivity change. Thus, the phase of the reflective electromagnetic wave is changing accordingly. In other words, we can modify the dielectric constant of the LC by changing the external voltage applied. However, the initial permittivity can only be recovered by removing the bias voltage and then relying on a polyimide (Pi) alignment layer in order to restore the initial orientation of the LC molecules. For instance, the reflective LC phase shifter that is proposed in [28,29] can produce a large phase shift by loading then external voltage for regulating the LC. However, the permittivity of the LC is restored to its initial state by a Pi-imposed alignment layer. Nevertheless, this approach inevitably brings some disadvantages. On the one hand, the Pi alignment layer after the friction process exhibits an unpredictable orientation, whereas different experimenters, different friction methods, or different batches of experiments may appear different friction orientations. This effect leads ultimately to a degree of uncertainty in the maximum dielectric constant, which further affects the performance of the LC-based device. On the other hand, the free relaxation time for the reorientation procedure of the LC molecules by the Pi alignment layer is too long, usually in seconds, minutes, or hours, which seriously influences the response rate of the phase shifter. Furthermore, the LC phase shifter proposed in [30,31] adopts full electronic control for regulating the LC and avoiding the above disadvantages. In terms of the former, only a 35° phase shift can be produced by enforcing a loading 100 V bias voltage at 1 THz. Obviously, the bias voltage is too large and the resulting phase shift is too small. Compared with the latter, the phase shifter proposed in this work adopts a multiple dipole structure, which not only enhances significantly the phase shift range but also renders it easier to adjust the bandwidth.
In summary, the conventional reflective phase shifter possesses many problems, since the LC free relaxation time is too long and there are difficulties with the fast and continuous modulation of the phase. Besides, the new fast modulation phase shifter based on LC also exhibits many defects, including the smaller phase shift range or the larger loading voltage.
In this work, a fully electrically controlled reflective phase shifter operating in the THz band with a metal-dielectric-metal structure based on nematic liquid crystal (NLC) is proposed. The employed structure is composed of a double dipole reflector array, metal grating electrodes, as well as an LC layer injected between them. On top of that, the metal grating electrodes are divided into two sets of comb grid electrodes, which are cross-distributed with each other. In addition, the electric field distribution in the LC layer is regulated by varying the driving voltage applied to the adjacent line gate in order to achieve a continuous modulation of the LC dielectric constant. Thus, the continuous manipulation of the phase of the electromagnetic wave is achieved. The acquired experimental outcomes demonstrate that the metal-dielectric-metal (MDM) structure exhibits a maximum phase shift of 249° at 103 GHz.
2. Principle and design
A schematic illustration of the 3D structure of the proposed fully electronically controlled reflective phase shifter based on NLC is disclosed in Fig. 1(a). It is interesting to notice that both the upper and lower parallel media substrates are composed of 480 um thick quartz separated by 45 um diameter polystyrene microspheres. Moreover, its relative permittivity and loss tangent are ɛr = 3.75 and tanδ = 0.002, respectively. The LC mixture (HFUT-HB01) is filled in the gap region between the upper and lower quartz dielectric substrate, whose dielectric constant ranges from ɛmin = 2.63 to ɛmax = 3.61 [32]. The metal microstrip patch was set on the lower surface of the upper quartz dielectric substrate, whereas its unit structure is divulged in Fig. 1(b). In unit cell, two-resonant elements based on parallel dipoles are used to increase the bandwidth of reflective phase shifter. As reported in [33], a numerical optimisation process had been implemented to obtain the given optimal value. A bias voltage line orthogonally connects the two dipole patches in order to form a patch unit, which acts as both a resonant structure, as well as a ground electrode. The period length of the metal patch unit was P = 950 µm, while the length and width of the two dipole patches were Lx1 = 128 µm, Ly1 = 640 µm, Lx2 = 128 µm and Ly2 = 684 µm respectively. Besides, the dipole patch pitch is D = 260 um, whereas the bias voltage line width is w = 13 um. On top of that, the upper surface of the lower quartz dielectric substrate is fully covered with a periodically arranged copper subwavelength metal grating structure, which consists of two sets of comb-based grid electrodes distributed crosswise to each other. Each comb grating cell contains the same array of the five gate electrodes, which are three constant voltage gratings and two variable voltage gratings, respectively. More specifically, the three constant voltage gratings are set on both sides and in the center of the unit, while the two variable voltage gratings are distributed between the three constant voltage gratings under the form of finger insertion, as is depicted in Fig. 1(c). Such a distribution of grating electrodes is intended to prevent interference from the electric fields between the adjacent units. Moreover, the line width of the metal grating is h = 20 um, whereas the distance is d = 170 um, which is employed not only as a reflective surface but also as a driving electrode.
Fig. 1. (a) Depiction of a fully electronically controlled reflective phase shifter structure. (b) Illustration of metal patch unit structure. (c) Schematic diagram of the metal grating electrode unit structure.
For the proposed device, firstly, the double dipole metal patch array face was grounded and +12 V DC voltage was applied to both the constant voltage grating, as well as the variable voltage grating in the initial state. At this time, the electric field in the LC layer was approximately distributed perpendicular to both the upper and lower dielectric substrates, as is demonstrated in Fig. 2(a). The orientation schematic of the LC is revealed in Fig. 2(c), as well as the dielectric constant of the LC is the maximum ɛmax. At that time, the voltage applied to the variable voltage grating was continuously reduced until the value of −12 V. At this moment, the LC layer produces an electric field that is distributed parallel to the upper and lower dielectric substrate, as is disclosed in Fig. 2(b). It is interesting to notice that the LC molecules are reoriented under the influence of the transverse electric field between the constant voltage grating and the variable voltage grating, whereas the orientation schematic is displayed in Fig. 2(d). During this orientation state, the dielectric constant of the LC is detected in the minimum value of ɛmin. By continuously modifying the voltage applied on the variable voltage grating between +12 V and −12 V, the LC dielectric constant can be regulated in order to change rapidly and continuously between ɛmax and ɛmin. With the constant change of the dielectric constant of the LC, the resonant frequency is altered, so that the fully electronically controlled continuous manipulation of the phase of the THz wave is implemented.
Fig. 2. Simulated static electric field strength in (a) initial state and (b) final state. Schematic diagram of the LCs’ orientation during the (c) initial state and (d) final state.
Owing to the insufficient coverage of the LC layer by the metal patch, the acquired outcome is an uneven distribution of the electric field within the LC layer. In order to evaluate the introduced phase error without considering neither anisotropy nor the inhomogeneity of the LC, a more accurate model based on the principle proposed in Ref. [34] was established. Along these lines, the employment of a partition simulation was carried out in order to divide the LC layer into several regions for achieving the optimum simulation results. More specifically, the LC dielectric constant was set to a constant value of ɛmax = 3.61 in the region around the metal grating electrode due to the manifestation of a strong electric field, which is located within 25 um around the grating. On top of that, the LC layer between the constant voltage grating and the variable voltage grating was set as a tunable region. The edge part was located around the unit, away from the metal patch, and thus the introduced electric field is weak and the LC dielectric constant is considered to be ɛmin = 2.63. Additionally, the specific partition model of the LC layer is divulged in Fig. 3(a). The performance of the LC phase shifter was analyzed by using the Finite Element Method (FEM). The employed theoretical model assumes normal incidence of an y-polarization plane wave on the structure (i.e., the $\vec{E}$ is parallel to the dipole’s long edge), while the phase shift unit adopts the periodic boundary condition in the x-axis and y-axis directions. We have to underline that as the LC dielectric constant in the tunable region decreases from 3.61 to 2.63, the resonant frequency is blue-shifted from the value of 101.7 GHz to 104.8 GHz, as is depicted in Fig. 3(b). The phase of the reflective wave at different dielectric constants is also simulated as is divulged in Fig. 3(c), where the maximum phase shift is 267° at 103.7 GHz.
Fig. 3. (a) Specific division model of liquid crystal (LC). The simulated results for the reflection coefficient at normal incidence versus frequency for different dielectric constant values (b) reflection loss (c) phase.
3. Experimental results and discussions
In order to verify the performance of the proposed tunable devices, we fabricated dipole patch arrays and metal grating electrodes by using a photolithography process (consisting of the vacuum copper plating, photoresist spin coating, exposure, development, and wet etching). As a result, a 30×30 size array of two dipole patch units and comb grating units was deposited on the inner surface of the quartz substrate, whereas the sample size was 4 ×4.5 cm2. The actual sample is illustrated in Fig. 4(a). Figure 4(b), 4(c) and 4(d) reveal the metallographic microscope photos of the double dipole patch, variable voltage grating, as well as the constant voltage grating, respectively. The employed measurement environment for the sample and the system is disclosed in Fig. 5(a) and 5(b). More specifically, Fig. 5(b) demonstrates the specific feeder method in the test. The constant voltage electrode was connected to constant voltage gratings of the test sample in order to provide constant biased voltage. Accordingly, the variable voltage electrode was coupled to variable voltage gratings of the test sample for delivering the variable voltage. Furthermore, the dipole reflection array surface is grounded. The spectral response of the sample is tested at room temperature by a vector network analyzer (Agilent N5224A), with an F-band module extender (N5262AW08) in combination with two horn antennas (the frequency range was 90–140 GHz, the 3 dB bandwidth was 12 degree, and the gain was 21 dB). The utilization of the dual-port measurement has many advantages, such as reduced coupling of the dual ports, fewer spurious effects, and in addition, dual-port measurement can compensate for environmental effects, thus exhibiting better measurement accuracy [35]. The dual output DC power supply provides the bias voltage, while the S21 of the vector network analyzer is the electromagnetic wave reflected from the sample. During the experiment, the sample was placed in the far-field position of the antenna. In order to reduce the influence of the surrounding environment, the measured sample was covered with polyurethane foam corner cones-based absorbing material.
Fig. 4. (a) Image of the fabricated prototype. Metallographic microscope photos of (b) double dipole patch, (c) variable voltage grating and (d) constant voltage grating.
Fig. 5. (a) Experimental test environment for the fabrication of the samples. (b) Illustration of the specific driving method.
The measured reflection loss at different voltages is presented in Fig. 6(a). From the acquired outcomes, it can be observed that by reducing the voltage loaded on the variable voltage grating from +12 V to −12 V the resonant frequency is first blue-shifted and then red-shifted by a small amount. More specifically, during the initial state, +12 V DC voltage was enforced on the variable voltage grating, while at this time, the resonant frequency was located at 102.5 GHz. Subsequently, the voltage applied on the variable voltage grating was continuously reduced to −4 V, and the resonant frequency was shifted right to 104.8 GHz. However, as the voltage applied to the variable voltage grating was further declined to −12 V, the resonant frequency was shifted left to 104.5 GHz. This effect was ascribed to the fact that when the voltage applied to the variable voltage grating was −4 V, this voltage has not yet allowed the LC above the variable voltage grating to be deflected. However, the adjacent grating electrodes produce a transverse electric field due to the manifestation of a positive and negative voltage difference, which makes the LC molecules located between them to be horizontally distributed, as is divulged in Fig. 6(d). Moreover, as the voltage applied to the variable voltage grating continues to be reduced down to the value of −12 V, in addition to the horizontal distribution of the LC molecules due to the transverse electric field generated between the adjacent gratings, the voltage also induces the LC located above the variable voltage grating to be vertically distributed, as is revealed in Fig. 2(d). Compared to the −4 V voltage, the LC dielectric constant increases, thus the resonant frequency is red-shifted to the left by a small amount afterward. On top of that, the phase shift curves at different voltages are disclosed in Fig. 6(b), while the phase shift exceeds the value of 180° in the range of 102.5 GHz–104.3 GHz. We have to underline that the maximum phase shift is 249° at 103 GHz.
Fig. 6. (a) Measured reflection loss of the metal-dielectric-metal (MDM) structure. (b) Measured phase of the MDM structure. (c) Variation of the phase versus the bias voltage at three different frequencies. (d) Schematic diagram of the LCs’ orientation when −4v DC voltage is applied to variable voltage grating.
Figure 6(c) illustrates the phase shift under the application of different bias voltages at three frequencies. In the simulation, the resonant frequency shifts from 101.7 GHz to 104.8 GHz. However the measurement shows the resonant frequency changed from 102.5 GHz (+12 V bias) to 104.8 GHz (−12 V bias), and the maximum phase shift is also reduced by 18°. The discrepancy of both the frequency and phase shift between the measurement and simulation is mainly attributed to the following reasons. Firstly, the manifestation of unavoidable dimensional errors in the manufacturing process could interpret this effect. Secondly, the test sample is consisted of a limited array of cells, while the simulations are performed in a periodic environment. Thus, not all the energy of the incident wave was reflected back, and the test results include direct transmission of energy. In addition, the dipole patch is not sufficient in order to completely cover the LC layer in the device, resulting hence in an uneven distribution of the electric field in the LC layer. Finally, the path, scattering, and receiving losses during the test are also potential reasons for the difference between the experimental data and the simulated outcomes that should not be ignored.
On top of that, Fig. 7 illustrates the phase shift as a function of the different applied voltages at the maximum phase shift point. During the initial state, +12 V is applied to both the constant voltage grating and the variable voltage grating. When the voltage enforced to the variable voltage grating is reduced below the value of +7 V, the phase shift can be stabilized above 180°.
Fig. 7. Distribution of the variation of the phase shift versus the bias voltage at 103 GHz. (The red line is the standard line with a phase shift of 180°)
4. Conclusions
A fully electronically controlled reflective phase shifter in the THz band based on the implementation of NLC is proposed in this work. This effect originates from the innovative replacement of the bottom metal reflective surface of the conventional reflective phase shifter with a metal grating electrode. Therefore, the full electronically controlled modulation of the electromagnetic wave phase was realized. Meanwhile, the orientation layer was omitted and as a result, the fabrication process was remarkably simplified. The acquired experimental outcomes demonstrate that the maximum phase shift of the proposed phase shifter is 249° at 103 GHz. The proposed LC-based phase shifter exhibits potential applications in the millimeter and terahertz reconfigurable reflectarrays.
Funding
National Natural Science Foundation of China (61871171); Guangxi Key Laboratory of Wireless Wideband Communication and Signal Processing, Guilin University of Electronic Technology (GXKL06200207).
Disclosures
The authors declare no conflicts of interest.
Data availability
Data underlying the results presented in this paper are not publicly available at this time but may be obtained from the authors upon reasonable request.
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