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Abstract
This paper reports a photonics-assisted joint radar and communication system for intelligent transportation based on an optoelectronic oscillator (OEO). By manipulating the optical multi-dimensional processing module inserted in the OEO loop, two phase-orthogonal integrated signals are generated with low phase noise and high frequency, as the communication data loaded on the overall polarity of radar pulses. At the receiver, single-channel matched filtering and two-channel IQ data fusion are utilized to retrieve the communication data and the range profile, without any performance deterioration of either. In this way, the contradiction between the performance of two functions existing in the previous scheme is solved, and the integrated performance can be further optimized as bandwidth increases. A proof-of-concept experiment with 2 GHz bandwidth at 24 GHz, which is the operating frequency of short-range automotive radar, is carried out to verify that the proposed system can meet the requirement of the intelligent vehicles in the short-range scene. A communication capacity of 335.6 Mbps, a range profile with a resolution of 0.075 m, and a peak-to-sidelobe ratio (PSLR) of 20 dB is demonstrated under the experimental condition. The error vector magnitude (EVM) curve and constellation diagrams versus received power are measured, where the EVM is −8 and −14.5 dB corresponding to a power of −14 and 6 dBm, respectively.
© 2021 Optical Society of America under the terms of the OSA Open Access Publishing Agreement
1. Introduction
The ever-growing interest in intelligent transportation makes it necessary for vehicles to fulfill the high-precision detection of the environment and the communication with accessible networks in the meantime [1–4]. The co-existence of communication and radar in the same system for cooperative work is becoming an essential desire to reduce power consumption and hardware redundancy [5]. Moreover, the development of millimeter-wave (mm-wave) communication technology [6] and the introduction of digital signal processing (DSP) in modern radar [7] give rise to a tending consistency of working frequency and hardware structure in radar and communication, which supports the feasibility of the integration. By assigning different frequency bands or time slots to each function, radar sensing and communication can be regarded to perform simultaneously [8,9]. However, spectrum congestion and a waste of resources are still unresolved. Dividing the phase control array into sub-arrays to separately emit radar and communication signals in different directions is another scheme to realize the co-existence of two functions [10]. Besides, a couple of linear frequency modulated wave (LFM) with opposite chirp rates were utilized in [11] as radar and communication signals for the dual-use system. But the radar and communication signals can’t be distinguished thoroughly in these methods, leading to mutual interference and deteriorate performance.
In contrast, sharing two functions within a single waveform is an effective solution with no interference and maximum efficiency [12,13]. In this case, the fusion and demultiplex of radar sensing and communication are realized in digital signal designing and processing. Limited by electronic bottlenecks, signals generated by digital to analog converter (DAC) cannot cover large bandwidth in high frequency, which affects the resolution of the radar. Frequency multiplication has a deterioration of phase noise, aggravating the communication quality and receiver sensitivity. The high loss and electromagnetic interference (EMI) of cable transmission is also a challenge for great geographical coverage in smart city networks.
Microwave photonics connects the world of electronics and optics, giving rise to much superiority such as high frequency, large bandwidth, and low EMI [14–16]. At present, many photonics-based technologies have shown preliminary potential in the integration of radar and communication [8,17–19]. Two functions are achieved by up-converting to different bands with a mode-locked laser (MLL) in [17], where magnanimous spectrum resources are occupied and wasted. In a photonics-based way, the classic radar waveform LFM is generated and modulated by amplitude shift keying (ASK) form to simultaneously deliver data [18]. But the additional ASK modulation inside the unity of radar pulse destroys the pulse compression results and the data rate is also limited, as a communication capacity of 100Mbps is achieved with the peak-to-sidelobe ratio (PSLR) of the range profile deteriorated from 14 dB to 9.5 dB. Orthogonal frequency division multiplexing (OFDM) signal is a typical communication waveform, from which radar information can also be extracted through the relative phase shift of different subcarriers as mentioned in [19]. But the radar sensing performance is poor with a range resolution of 30 m. And the high peak to average ratio (PAPR) of OFDM, which is vulnerable to nonlinear distortions, makes it difficult to operate at high power for long-distance transmission. These results indicate that there is a trade-off between the performance of radar and communication, leading to the failure of two functions to be optimized at the same time. Moreover, an external microwave source is necessary for these methods, which results in poor phase noise and increasing complexity of the system. The Optoelectronic oscillator (OEO) is an effective solution to generate signals based on photonics [20]. It can generate and self-sustain wideband tunable, low phase-noise single frequency signal in both electrical and optical domain, with the independence of phase noise and frequency. Taking advantage of the benefits, multi-functional signals have been further produced for diverse applications with structural modifications of the OEO [21–25]. For example, the multi-frequency signal for multi-band radar [21], the chaotic signal for secure communication [22], the Fourier domain mode-locked signal for chirped radar [23], and the vector signal for radio-over-fiber (RoF) systems have been generated based on OEO [24] in recent years. Though the above research has shown the potential of OEO, the joint radar and communication system has not been reported yet with the bifunctional management of radar and communication in the OEO loop remaining to be solved.
In this paper, a photonics-assisted joint radar and communication system based on OEO is proposed and experimentally demonstrated. With the multi-dimensional processing module inserted, the light in the OEO can be polarization multiplexed and separately manipulated. In X-pol, a radio-frequency (RF) carrier of high frequency and low phase noise is sustained, inducing a frequency-shifted optical sideband. In Y-pol, in-phase and quadrature (IQ) amplitude modulation is utilized for two-channel radar-communication integration with large bandwidth and flexibility in optical baseband. The RF joint signal is generated through a balanced detection, with communication data loaded on the overall polarity of the binary-coded radar signals. By the IQ receiving and separate matched filtering, two pulse trains are obtained with different positions and polarity. Utilizing the correlated fusion of two channels breaks the joint performance limit and eliminates the radar range ambiguity caused by high-speed communication in single channel. And along with the increase of the bandwidth, the joint performance can be further optimized. Thus, the communication data and range profile can be retrieved with independent and promising performance. A radar range profile with a resolution of 0.075 m, a PSLR of 20 dB, and a communication capacity of 335.6 Mbps is experimentally demonstrated under the condition of 2 GHz bandwidth and 24 GHz center frequency. The EVM curve versus received power are also measured to verify the communication quality. When the power is −14 and 6 dBm, the EVM is −8 and −14.5 dB respectively and the corresponding constellation diagrams are also shown.
2. Principle
The schematic diagram of the proposed photonics-assisted joint radar and communication system based on OEO is shown in Fig. 1(a). The lightwave emitted from a laser diode (LD) passes through a multi-dimensional processing module, which can carefully control the amplitude, phase, frequency, and polarization direction as illustrated in Fig. 1(b).
Fig. 1. (a) Schematic diagram of the proposed photonics-assisted joint system. (b) Detailed structure of the optical multi-dimensional processing module. EC: electrical coupler; OC: optical coupler; BPF: bandpass filter; EA: electrical amplifier; PBS: polarization beam splitter; PBC: polarization beam combiner; PC: polarization controller; SMF: single-mode fiber; PD: photodetector; BPD: balanced photodetector; DSP: digital signal processing.
In the processing module, the incident light is divided into two parts by polarization multiplexing. The frequency modulation in X-pol is a self-sustaining process. Part of the output of the processing module passes through a section of single-mode fiber (SMF), a photodetector (PD), an electrical amplifier (EA) to compensate for the loop loss, a bandpass filter (BPF) to select the mode, and is finally fed back to the processing module, exactly as an OEO. When the self-excited oscillation starts, an electrical single frequency signal is maintained and an optical frequency-shifted sideband is generated as well. However, the optical carrier also needs to be retained to beat with the sideband for the reproduction of the electrical signal in OEO. The optical output in X-pol can be expressed as:
where ${E_c}$ and ${\omega _c}$ are the amplitude and angular frequency of the input light, ${\omega _e}$ is the angular frequency of the oscillation signal in OEO, which depends on the center frequency of the BPF. In Y-pol, the optical amplitude and phase are manipulated with high speed and large bandwidth to realize the integration of radar and communication in baseband. Considering two general pulse compression radar sequences ${r_I}[n]$ and ${r_Q}[n]$, with communication data ${c_I}$ and ${c_Q}$ modulated on the overall polarity, dual-functional joint sequences $r{c_I}[n]$ and $r{c_Q}[n]$ can be generated as:Fig. 2. (a) Spectrum evolution process in the proposed joint system. (b) Principle of integrated demodulation of radar and communication
The RF integrated signal is sent to the free space to interact with the environment. It can be reflected by obstacles for anti-collision warning, and can also be received by other vehicles for data transmission. In the meantime, other vehicles may transmit integrated signals as well.
A signal extracted from the OEO helps to orthogonally down-convert the echo ${f_1}(t - \Delta t)$ at the local receiver. After the frequency mixing and low pass filter, ${f_{2I}}(t)$ and ${f_{2Q}}(t)$ are obtained:
In a single channel, a detected distance of $c\Delta {t_I}/2$, a range resolution of $c\tau /2 = c/2B$, a maximum unambiguous range ${R_{\max }}$ of $Mc\tau /2$, and a communication capacity C of $1/M\tau $ is obtained. It can be seen that there is a contradiction between the C and ${R_{\max }}$ owing to a fixed product of the two: $C \cdot {R_{\max }} = c/2$ as the joint performance limit. We can only improve ${R_{\max }}$ by increasing M, resulting in a significantly poor C. It’s the same for the other channel, just with a replacement of I to Q and M to N. To solve this problem, two individual channels are fused by DSP as illustrated in Fig. 2(b). By projecting two channels to an identical dimension, the communication data is completely preserved and superimposed, leading to an expanded capacity of $1/M\tau + 1/N\tau $. For the range profile, the double confidence principle of real target invariance of is applied, where the real distance $r = c\Delta t/2$ is obtained only when it simultaneously satisfies $\Delta t\textrm{ = }\Delta {t_I} + i \cdot M\tau = \Delta {t_Q} + j \cdot N\tau $, where i and j are positive integers. Since M and N are coprime, the maximum unambitious range is increased to $MNc\tau /2$. The joint performance is $C \cdot {R_{\max }} = (M + N) \cdot c/2$, with the limit of a fixed product been broken. Utilizing another channel with a large N which is coprime with M to increase the radar performance ${R_{\max }}$, leading to a slight improvement rather than a significant deterioration in the communication performance C. According to this idea, two functions may perform well independently by skillful adjustment of the parameters.
3. Experimental setup and results
Figure 3 shows the proof-of-concept experimental setup of the proposed system. The polarization-multiplexing dual-parallel Mach-Zehnder modulator (PM-DPMZM) is an integrated device possessing an IQ amplitude modulator in each polarization, which plays an important role in the optical multi-dimensional processing module. The light wave from the LD with a wavelength of 1550 nm and a power of 14 dBm is injected into the PM-DPMZM (FTM7980EDA/301). Amplified by an erbium-doped fiber amplifier (EDFA) to 13 dBm, the output of the PM-DPMZM is divided into two parts by a 50:50 optical coupler (OC). One passes through an SMF of 4 km length, a PD (u2t, XPDV2120RA), an EA with 40 dB gain, a BPF with a center frequency of 24 GHz and 50 MHz bandwidth, a 90° hybrid, and then feeds back to the DPMZM in X-pol.
Fig. 3. Experimental setup for the proposed photonics-assisted joint system. PM-DPMZM: polarization-multiplexing dual-parallel Mach-Zehnder modulator; HPF: high-pass filter; AWG: arbitrary waveform generator; OSA: optical spectrum analyzer; ESA: electrical spectrum analyzer; OSC: oscilloscope;
Two outputs of the 90° hybrid are applied to two arms of the DPMZM separately with three quadrature biases. Thus, a self-excited oscillation at 24 GHz with low phase noise is sustained in OEO and the optical single sideband (SSB) modulation is obtained in X-pol, as the blue line of Fig. 4(a) shows. The phase noise measured by a phase noise analyzer (Agilent N9030A) is presented in Fig. 4(b), which reaches −123 dBc/Hz at 10 kHz offset. In Y-pol, two baseband integrated signals from the arbitrary waveform generator (AWG, Tektronix AWG70002A) are applied to two arms of the other DPMZM for IQ amplitude modulation with the phase biases ${\varphi _{B1}} = {\varphi _{B2}} = {\varphi _{B3}} = \pi /2$. Two radar signals are individually encoded by 11-bit and 13-bit M-sequence with a bandwidth of 2 GHz to obtain the pulse compression characteristics. The random communication data is loaded on the overall polarity of radar pulses whose duration is 5.5 ns and 6.5 ns respectively. The optical output of Y-pol measured by an optical spectrum analyzer (OSA) is shown as the red line in Fig. 4(a).
Fig. 4. (a) Optical spectrum of x-pol and y-pol. (b) Phase noise of OEO. (c) Electrical spectrum of generated joint signal. (d) Amplitude and phase of generated two-channel joint signal.
The other output of the OC which contains two orthogonal polarized lights is injected into the PBS through a PC. After a BPD (u2t, BPRV2125AM), a high pass filter (HPF) with a passband of 18∼26 GHz, and an EA with 35 dB gain, the RF dual-functional integrated signal is generated and measured by a spectrum analyzer (ESA, Agilent E4446) as shown in Fig. 4(c). Adjusting the PC until the carrier component becomes minimum, the balanced detection is realized and the baseband dual-functional operation is converted to RF through the OEO with maximum efficiency. The center frequency is 24 GHz which is the same as the OEO, and the bandwidth is 2 GHz which is consistent with the baseband joint signal. By an external local oscillator (LO) with ${f_{LO}}$=12 GHz, the RF integrated signal is down-converted to intermediate frequency (IF) with ${f_{IF}} = {f_e} - {f_{LO}}$=12 GHz which is within the bandwidth of the digital oscilloscope (OSC, Keysight UXR0134A). The down-converted signal ${f_{IF}}(t)$ is:
The time-domain waveform of ${f_{IF}}(t)$ is collected by the OSC and the phase information is demodulated by Hilbert transformation in Fig. 4(d). The four-step phase change illustrates clearly that the generated RF signal at transmitter is the sum of two orthogonal channels with a relative phase difference of $\pi /2$.
At the receiver, an IQ mixer is utilized to demodulate two channels of integrated signals in the baseband, with the help of a local oscillator which is extracted from the OEO through an electrical coupler (EC). In the laboratory, a direct connection between transceivers substitutes for wireless transmission as a back-to-back transmission experiment to verify the dual function of the proposed system. Two outputs of the IQ mixer ${f_{2I,Q}}(t)$ are collected by the OSC as shown in Fig. 5(a) and Fig. 5(b). It can be seen that the function integration is fulfilled as the communication data ${c_{I,Q}}$ is loaded on the overall polarity of two radar pulses ${r_{I,Q}}(t)$ with different coding sequences ${r_{I,Q}}[n]$ and coding lengths M and N. The following DSP is implemented to retrieve the communication information and the radar range profile. By cyclic cross-correlating with ideal radar signals ${r_I}(t)$ and ${r_Q}(t)$, the waveform of each channel is compressed into a series of pulses with different polarity as ${f_{3I,Q}}(t)$ in Fig. 5(c) and Fig. 5(d).
Fig. 5. (a) Received baseband joint signal of I-channel. (b) Received baseband joint signal of Q-channel. (c) Pulse compression result of I-channel. (d) Pulse compression result of Q-channel.
The relative position of each pulse in one period represents the transmission delay while the pulse polarity carries one bit of communication data. In each channel, one period of pulses is intercepted whose square value is shown in Fig. 6(a) and Fig. 6(b) as the radar range profile. The range resolution is 0.075 m, corresponding to the bandwidth of 2 GHz. The duration of the in-phase and quadrature channels is 5.5 ns and 6.5 ns, leading to a maximum unambiguous range of 0.825 m and 0.975 m respectively. The PSLR is 20.8 and 19 dB respectively, depending on the autocorrelation performance of the adopted M-sequences, which indicates the loaded data make no impact on the radar performance. The detected distance indicated by the pulse position of an individual channel is periodicity ambiguity and unreal. According to the double confidence principle of real target invariance which has been explained in the previous chapter, a digital fusion is implemented as shown in Fig. 6(c) with the maximum unambiguous range expanded to 10.725 m and a PSLR of 20 dB.
Fig. 6. (a) Radar range profile of I-channel. (b) Radar range profile of Q-channel. (c) Radar range profile by fusing I-channel and Q-channel.
The communication information ${c_{I,Q}}$ is obtained from the peak value of ${f_{3I,Q}}(t) = {c_{I,Q}} \cdot {R_{I,Q}}(t - \Delta {t_{I,Q}})$ in Fig. 5(c) and Fig. 5(d). Due to the phase jitter and other noises in the system, the normalized peak values have fluctuations around ${\pm} 1$. To measure this fluctuation, 1000 normalized pulse peak values of I and Q-channel are collected as the horizontal and longitudinal ordinates respectively, and the constellation is obtained for the intuitive expression of communication quality. For a more accurate description, EVM is also calculated according to the formula where ${c_{I,Q}}$ are the ideal information and ${c_{I,Q}}^{\prime}$ are the demodulated information:
By adding an optical attenuator in front of the BPD, the EVM is measured as a function of received power, which is illustrated in Fig. 7 and the constellations with received power of −14 dBm and 6 dBm are showed in the insert. The communication capacity of I and Q channel is 181.8 and 153.8 Mbps separately. The total communication capacity is the sum of the above, which is 335.6 Mbps.
Fig. 7. The EVM curve via received power. Insert: the constellation diagram when power is −14 dBm and 6 dBm.
4. Discussion
The performance of the proposed joint radar and communication system, which is range resolution $\Delta R$, maximum unambiguous range ${R_{\max }}$, and communication capacity C, is relevant to the bandwidth B and the radar coding length M, N as shown in Table 1.
Table 1. Relationship between integrated radar and communication performance and system parameters
The theoretical ${R_{\max }}$ and C of single-channel and IQ dual channels with different B are calculated in Fig. 8 according to Table 1. Under the same bandwidth of 2 GHz and a consequent $\Delta R$ of 0.075 m, ${R_{\max }} \cdot C$ is small and fixed in an individual channel leading to a poor joint performance, as the red and blue line in Fig. 8. By adding another channel and digital fusion under the same bandwidth, ${R_{\max }} \cdot C$ is greatly enlarged and not a constant anymore, as the yellow line in Fig. 8. If the bandwidth is expanded to 4 GHz and $\Delta R$ becomes 0.0375 m, ${R_{\max }}$ can be further doubled while $C$ remains the same, as the purple line in Fig. 8. Nevertheless, under a fixed bandwidth with IQ channels, there is still a negative relationship between ${R_{\max }}$ and $C$ according to different options of M and N. For the short-range automotive radar working at 24 GHz, the main applications are blind-zone monitoring, parking assistant, and other early warning functions, whose detection range is 1∼100 m depending on different scenarios. Under the experimental condition in this paper (B=2 GHz, M=11, N=13), the results of $\Delta R$=0.075 m, ${R_{\max }}$=10.725 m, and C=335.6 Mbps are demonstrated to be consistent with the theory and are in accord with the demand of short-range radars. In specific applications, the parameters may be selected carefully for a larger ${R_{\max }}$ and a trade-off in C according to the rule illustrated in Fig. 8.
Fig. 8. Integrated radar and communication performance of single-channel and IQ dual channels with different bandwidth.
There is another significant problem in practical application owing to OEO. Due to the use of long optical fiber, OEO has the characteristics of a promising short-term phase stability but a poor long-term frequency stability. Firstly, the fiber length is sensitive to ambient temperature and vibration which result in a continuous frequency drift. Secondly, the narrow mode spacing of OEO leads to the discrete mode hopping within the bandwidth of the filter. If the frequency of transmitter and receiver cannot keep consistent and synchronous, the baseband demodulation of radar and communication cannot be completed. Many methods have been proposed for the long-term stability of OEO frequency, such as employing environmental-insensitive fiber [26], injection-locking [27], phase and delay compensation by feedback control [28,29]. The Allen deviation has reached 4*10–11 at an averaging time of 10000 s, which is the same order of magnitude as commercial oscillator (Keysight E8257D). In this manuscript, a proof-of-concept back-to-back experiment is carried out to verify the validity of the core principle. The same OEO is used in both transmitter and receiver and the frequency drift is eliminated with no effect on the principle verification. In real application scenarios, the OEO frequency can be stabilized under the guidance of the above methods to meet the need of demodulation of radar and communication.
5. Conclusion
In conclusion, we have proposed and demonstrated a photonics-assisted joint radar and communication system based on an OEO. By modulating the overall polarity of radar signals with binary communication data, the dual-functional joint signal is generated in the baseband. With an optical multi-dimensional processing module inserted in the OEO, the polarization, phase, amplitude, and frequency of the light can be manipulated carefully by the baseband joint signal and the RF carrier. In this way, two channels of RF dual-functional joint signal with orthogonal phase are directly produced without an external microwave source. At the receiver, the range profile and communication data can be obtained by IQ matched filtering of individual channel and the digital fusion of two channels. The joint performance limit in single channel is broken with the range ambiguity caused by high-speed communication eliminated, making it possible to independently obtain promising radar and communication characteristics. A back-to-back proof-of-concept experiment was carried out to verify the performance of the proposed system. Under a bandwidth of 2 GHz and a center frequency at 24 GHz, a range profile with a resolution of 0.075 m, a PSLR of 20 dB, a maximum unambiguous range of 10.725 m, and a communication capacity of 335.6 Mbps is experimentally demonstrated. The EVM curve via received power and the corresponding constellation diagrams are also obtained, among which the EVM is −8 and −14.5 dB when the power is −14 and 6 dBm. The influence factor and possibility for further optimization of the performance with IQ channels are discussed, according to which one can the adjust the experimental parameters for a specific scenario in intelligent transportation.
Funding
National Key Research and Development Program of China (2019YFB2203301); National Natural Science Foundation of China (61690191).
Disclosures
The authors declare no conflicts of interest.
Data availability
No data were generated or analyzed in the presented research.
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