Anti-dispersion phase-tunable microwave mixer based on a dual-drive dual-parallel Mach-Zehnder modulator

Mutong Xie; Mingyang Zhao; Mingzheng Lei; Yongle Wu; Yuanan Liu; Xinlu Gao; Shanguo Huang

doi:10.1364/OE.26.000454

1. Introduction

Modern radar is widely used in target acquisition, missile guidance, detection and search, air traffic control, weather-sensing, etc [1,2]. Phase shift is a key technology in the phased array antenna (PAA), which is widely adopted in modern radar systems, to improve the detection and tracking ability. This is because electronic scanning provides rapid beam steering, compared to the traditional mechanically scanned antenna system. Moreover, as military radar develops, frequency hopping emerges as an outstanding method to combat deliberate jamming. Thanks to the distinctive advantages of the microwave photonics technology, such as small footprint, large bandwidth, constant attenuation over the entire microwave frequency range [3] and immunity to electromagnetic interference, it is of great interests to implement optically-controlled frequency mixing and phase-shifting in phased array antenna systems.

To reduce bulkiness and improve the overall performance, a frequency mixer and a phase shifter can be combined into a simple compact configuration, which is an optically-controlled phase-tunable microwave mixer [4–6]. Such a mixer can be realized through cascaded phase modulators [7], dual-parallel Mach-Zehnder (MZ) modulators [8], polarization modulators [9], and Hilbert transform path [10]. The scheme of cascaded modulators structure has high isolation between the local oscillator (LO) port and the microwave input port. The parallel structures using MZ/polarization modulators have higher power efficiency, and often require an optical band-pass filter (such as an optical grating) to filter out different sidebands with different frequencies. In the scheme based on Hilbert transform path [10], phase-tunable mixing is achieved without using any optical filters, which reduces system complexity and improves robustness.

In a modern radar network with multistatic structure, fiber transmission is a practical way to connect radar stations because of high speed and low interference. However, the phase-tunable microwave mixer meets the problem of frequency-selective power fading due to the chromatic dispersion in long-range fiber transmission [11, 12]. It can be solved by either introducing an extra dispersion compensation fiber (DCF) with opposite dispersion [13], or performing optical conjugation in the middle of the link [14], or tuning the phase of each frequency component by an optical filter [15]. Those schemes require additional optical devices in the link, resulting in low compensation flexibility. Schemes based on optical-electrical modulator have been studied for higher flexibility and lower bulkiness [16–19]. For narrowband signals, double side-band modulation with phase-shifted carrier results in a simple structure with good performance in chromatic dispersion compensation [16, 17], but the compensation error rises when the input signal bandwidth increases. For wideband signals, Hilbert transform path method makes the system immune to chromatic dispersion power fading [18, 19], regardless of signal bandwidth. However, it is hard to implement the Hilbert transform path because the phase difference between the LO and the input signal are only concerned with chromatic dispersion, and cannot be altered by changing the mixer configuration. Therefore, it is challenging to realize wideband dispersion-free phase-tunable mixer without the use of optical filters.

This paper proposes a wideband dispersion-free optically-controlled phase-tunable mixer scheme based on a dual-drive dual-parallel modulation Mach-Zehnder modulator (DDDP-MZM). Phase-tunable single side-band modulation for the LO signal and double side modulation for the input signal is used. Firstly, the structure avoids vector superposition for the same output microwave frequency, thus eliminating the dispersion-related frequency-selective fading. Secondly, direct electrical tuning of the wideband microwave input signal is avoided, thus supporting large working bandwidth. To improve system robustness and lower system complexity, no optical filter is used in the system. The rest of the paper is organized as follows: section 2 provides the system principle and structure, section 3 analyzes the experimental records, and section 4 gives the conclusion.

2. Theory and system structure

In a typical optically-controlled frequency mixer system, the input signal (RF/IF) and LO signal are modulated onto the optical carrier, and frequency mixing is done by beating these modulated signals. To consider using conventional intensity modulation for signals and ignoring the suppressed optical carrier, the modulated signal can be expressed as

E_{M} = \sqrt{\frac{P_{o}}{8}} \exp (j ω_{o} t) [2 m_{i n} \cos (ω_{i n} t) + 2 m_{L O} \cos (ω_{L O} t)]

where

P_{o}

is the power of the optical carrier,

ω_{o}

,

ω_{i n}

and

ω_{L O}

are the angular frequency of the optical carrier, the input signal and the LO signal, respectively.

m_{i n}

and

m_{L O}

are the modulation index of the input signal and the LO signal, respectively. If the second-order chromatic dispersion is considered, the signal after fiber transmission can be written as

\begin{matrix} E_{M}^{'} = \sqrt{\frac{P_{o}}{8}} \exp (j ω_{0} t) [2 m_{i n} \cos (ω_{i n} t) \cos (\frac{β_{2} L ω_{i n}^{2}}{2}) \\ + 2 m_{L O} \cos (ω_{L O} t) \cos (\frac{β_{2} L ω_{L O}^{2}}{2})] \end{matrix}

where

β_{2}

is the second-order chromatic dispersion parameter in frequency domain, and

L

is the transmission distance. Then, the beating signal after photodetector (PD) is (optical carrier is suppressed)

I^{'} = \frac{1}{4} P_{o} R A_{i n} A_{L O} {\cos [(ω_{L O} - ω_{i n}) t] + \cos [(ω_{L O} + ω_{i n}) t]} \cdot \cos φ_{2}

where

φ_{2} = \frac{β_{2} L}{2} (ω_{L O}^{2} - ω_{i n}^{2})

is the frequency-selective power fading term, which results in periodical power notches in the microwave spectrum. To mitigate this fading term and tune the phase of the output mixed signal, we propose to perform single side-band modulation for the LO signal, and electronically shift the phase of the LO signal, which is

\begin{matrix} E_{M}^{' ​'} = \sqrt{\frac{P_{o}}{8}} \exp (j ω_{0} t) [2 A_{i n} \cos (ω_{i n} t) \cos (\frac{β_{2} L ω_{i n}^{2}}{2}) \\ + A_{L O} \exp (ω_{L O} t + φ) \cos (\frac{β_{2} L ω_{L O}^{2}}{2})] \end{matrix}

where

φ

is the phase shift of the LO signal. Since the LO signal is side-band modulated, beating of the input and the LO signal would not cause a vector superposition between double side-bands like DSB modulation, and LO’s phase shift can be transferred to the output signal. Thus, the output current is

\begin{matrix} I^{″} = \frac{1}{8} P_{o} R A_{i n} A_{L O} {\cos [(ω_{L O} - ω_{i n}) t + \frac{β_{2} L}{2} (ω_{L O}^{2} - ω_{i n}^{2}) + φ] \\ + \cos [(ω_{L O} + ω_{i n}) t + \frac{β_{2} L}{2} (ω_{L O}^{2} - ω_{i n}^{2}) + φ]} \end{matrix}

From (5) it can be seen that the second-order chromatic dispersion term remains a phase term rather than being transformed into an amplitude term, and the power fading is eliminated. Meanwhile, the phase shift of the output signal is achieved by processing the single frequency LO signal, avoiding directly processing the wideband input signal with electrical devices, making the scheme wideband-friendly. Figure 1 shows proposed scheme structure. The main component of the scheme is the dual-drive dual-parallel Mach-Zehnder modulator (DDDP-MZM). Two identical MZs, MZ1 and MZ2 are integrated on the two arms of the DDDP-MZM, and the two arms form another MZ, which is called MZP. An optical carrier is sent to the DDDP-MZM. The LO signal is separated by a 1x2 power splitter, after going through a variable electronic phase shifter to acquire an initial phase shift $φ$ . Then, a phase shifter is used in one of the routes to create a 90° phase difference between the two routes, and these two orthogonal routes are sent to the two arms of MZ1, respectively. The bias voltage at MZ1 is $V_{π} / 2$ , to gain a 90° phase difference between the optical carriers at each arm. Thus, the output of MZ1 is (ignoring the high-order terms)

Fig. 1 System structure of the anti-dispersion phase-tunable microwave mixer based on DDDP-MZM. (VSG: Vector signal generator. PS: Power splitter. MZ: Mach-Zehnder modulator. OC: Optical coupler. LD: Laser diode. VNA: Vector network analyzer. OSA: Optical spectrum analyzer. MSA: Microwave spectrum analyzer. SMF: Single mode fiber. EDFA: Erbium-doped fiber amplifier. PD: Photodiode. DDDP-MZM: Dual-drive dual-parallel Mach-Zehnder modulator.)

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E_{o u t 1} = \sqrt{\frac{P_{o}}{16}} {j 2 J_{1} (m_{L O}) \exp [j (ω_{o} - ω_{L O}) t - j φ] + (1 + j) J_{0} (m_{L O}) \exp (j ω_{o} t)}

From (6) it can be seen that the output of MZ1 is a phase-tuned single side-band modulation for the LO signal. At MZ2, the input signal (IF/RF) is only sent to one of its arms, and the output is

\begin{matrix} E_{o u t 2} = \sqrt{\frac{P_{o}}{16}} {j J_{1} (m_{i n}) \exp [j (ω_{0} - ω_{i n}) t] + (j + J_{0} (m_{i n})) \exp (j ω_{o} t) \\ + j J_{1} (m_{i n}) \exp [j (ω_{0} + ω_{i n}) t]} \end{matrix}

The two output of MZ1 and MZ2 are combined at MZP, and the bias voltage is set to $V_{π}$ , to make the optical carriers of MZ1 and MZ2 eliminate each other. If $m_{i n} = m_{L O} = m$ , and the small signal approximation $J_{n} (z) \approx z^{n} / (n - 1)!$ is used, the output becomes

\begin{matrix} E_{o u t} = j \sqrt{\frac{P_{o}}{16}} {J_{1} (m) \exp [j (ω_{o} - ω_{L O}) t - j φ] + A_{e r r} \exp (j ω_{o} t) \\ - J_{1} (m) \exp [j (ω_{0} - ω_{i n}) t] - J_{1} (m) \exp [j (ω_{0} + ω_{i n}) t]} \end{matrix}

where

A_{e r r} = J_{0} (m) - 1

is the small signal approximation error, which indicates the vestigial carrier that is not completely eliminated. Thus, the beating signal from the PD is

I_{o u t} = I_{C 1} + I_{C 2} + I_{e r r}

where

I_{C 1}

and

I_{C 2}

are the output signal of down-conversion and up-conversion, respectively. They are resulted from the beating of the LO signal and the two side-bands of the input signal, which are

I_{C 1} = \frac{P_{o}}{16} R J_{1}^{2} (m) \cos [(ω_{L O} - ω_{i n}) t + φ]

I_{C 2} = \frac{P_{o}}{16} R J_{1}^{2} (m) \cos [(ω_{L O} + ω_{i n}) t + φ]

and

I_{e r r}

is the beating from vestigial carrier and other side-bands, which is

I_{e r r} = \frac{P_{o}}{16} R J_{1} (m) A_{e r r} [\cos (ω_{L O} t + φ) - 2 \cos (ω_{i n} t)]

Thus, wideband phase-tunable optically-controlled frequency mixing is realized by this scheme, which simultaneously generates the up-conversion and down-conversion signal, and is immune to the power fading caused by chromatic dispersion.

3. Experiment results

An experiment is conducted to test the frequency mixing performance of the system, which uses the structure shown in Fig. 2. In this experiment, the DDDP-MZM used is Fujitsu FTM7960EX (with PlugTech MBC-IQ-03 bias controller attached), the wavelength of the laser source is 1550.561 nm, a vector network analyzer (VNA) (Agilent B722ES) is used to generate input (RF/IF) microwave signal, a vector signal generator (VSG) (Agilent E8257D) is used to generate the LO signal, and an EDFA is added before the PD for higher output power. The extinction ratio of the MZM is measured to be 59 dBm without the EDFA, and 36 dBm when the bias current of the EDFA is 133 mA, due to the rise of the optical noise floor. When the input signal $f_{i n}$ is set at 14 GHz, and $f_{L O}$ is 18 GHz, the output optical spectrum is shown in Fig. 3(a) and the beating microwave signal spectrum (between 1 GHz and 19 GHz) is shown in Fig. 3(b). In Fig. 3(a) the power of the vestigial upper (in terms of frequency rather than wavelength) LO side-band is −22.7 dBm. Compared to the remaining lower side-band which has a power of 0.448 dBm, the rejection ratio is 23.1 dBm. The vestigial carrier signal is also kept at a relatively low power level, as well as the second-order harmony frequencies. Figure 3(b) is acquired with a microwave spectrum analyzer (Agilent MXA N9020A). In Fig. 3 (b) the output down-conversion signal has a power of −26.43 dBm, and almost no LO or input signals are observed. Figure 3(c) shows the optical spectrum when the input signal is 4 GHz and the LO signal is 8 GHz. The beating output is shown in Fig. 3(d), where the power of up-conversion output is −27.87 dBm, and the power of down-conversion output is −25.23 dBm. There also exists a double frequency signal (−25.87 dBm). The three signals can be further separated and selected in microwave domain.

Fig. 2 Experiment structure of the anti-dispersion phase-tunable microwave mixer based on DDDP-MZM to test the frequency mixing performance.

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Fig. 3 (a) The optical spectrum and (b) the beating output when input signal is 14 GHz and the LO is 18 GHz. (c) The optical spectrum and (d) the beating output when input signal is 4 GHz and the LO is 8 GHz.

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To investigate the phase-shift performance of the system, an electrical mixer (Marki M1R0726) is used to reconvert the output signal back to the input frequency, and the VNA is used to observe and record the S21 power and phase parameters. The experimental structure is shown in Fig. 4(a), and the “transmission block” here contains only an EDFA, as is shown in Fig. 4(b). The port 1 of the VNA is connected to the input of the mixing module, and the amplified output of the mixing module, is sent to a PD before mixing with the LO signal for reconversion. The electrical mixer has a frequency range of DC ~8 GHz for the IF port, and 7 ~26.5 GHz for the RF and LO port. Based on this frequency limitation, the measurements are taken over the frequency between 12 ~16 GHz, and the LO signal is set as 18 GHz. The down-conversion signal is passed to the VNA, while the up-conversion signal with a frequency range of 30 ~34 GHz, which exceeds the up limit of the electrical mixer, is rejected. Thus, no additional electrical filter is used in the testing system. The VNA is calibrated in through mode, and the initial phase value is adjusted stepwise by 30°, so that the phases of the output signal is shifted stepwise by 30°. The S21 power and phase response at different initial phase-shift $φ$ are shown in Fig. 5. From Fig. 5(a) and 5(b) it can be seen that a 0 ~390° tunable flat-phase and flat-power responses are achieved, with the highest phase deviation less than 5° and power variation less than 3.5 dB. Figure 5(c) and 5(d) shows the long-term stability of the mixer, where the input signal is 14 GHz and the sweep time is 200 s. The long-term max phase variation is only 4.48°, and max power variation is only 0.84 dB. Figure 6 shows the dynamic range performance. The noise floor is −150 dBm/Hz, and the spur free dynamic range (SFDR) is measured to be 88.26 dB·Hz^2/3.

Fig. 4 (a) Experiment structure of the anti-dispersion phase-tunable microwave mixer based on DDDP-MZM with different transmission process, (b) to test the phase-shift performance and (c) to test the anti-dispersion performance.

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Fig. 5 The (a) phase and (b) power response at different initial phase $φ$ when the input signal sweeps between 12 ~16 GHz. The LO signal is 18 GHz. The long-term (200 s) (c) phase stability and (d) power stability (compared to mean power) of the proposed scheme at 14 GHz.

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Fig. 6 Spur-free dynamic range performance of the proposed scheme at 14 GHz. (IMD3: third-order intermodulation distortion).

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To test anti-dispersion performance of the mixer, a 51 km SMF is added in the transmission process, as is shown in Fig. 4(c). Figure 7 shows the measured output power as the function of input frequency with CS-DSB (shown in red circles) or CS-SSB (shown in blue asterisks) modulation schemes for the LO signal. The measurement interval is 0.2 GHz. When CS-DSB modulation is adopted for the LO signal, 2 power notches caused by chromatic dispersion at 12.2 GHz and 17.2 GHz can be seen, while no notches are seen when CS-SSB modulation for the LO signal is adopted, proving that the CS-SSB modulation for the LO signal used in the proposed scheme eliminates the power fading brought by second-order chromatic dispersion.

Fig. 7 Measured output power as the function of the input frequency with different modulation schemes for the LO signal.

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4. Conclusion

An optical anti-dispersion phase-tunable microwave mixer is proposed and experimentally demonstrated. Based on DDDP-MZM, the anti-dispersion and phase-tunable function without the need for optical filters is achieved by introducing phase-tunable single side-band modulation for the LO and double side modulation for the input signal. The mixer is suitable for wideband signal because the phase shift of the output signal is achieved by processing the single frequency LO signal, and directly tuning the wideband input signal is avoided. Experiments are conducted to test the frequency mixing ability, the phase tuning ability and the anti-dispersion performance, showing a phase variation of less than 5°, a power variation less than 3.5 dBm, and no power notches despite 51 km SMF transmission. In conclusion, proposed scheme is suitable for optically-controlled time-delay/phase-shift antenna array beamforming networks.

Funding

National Natural Science Foundation of China (NSFC) (61690195, 61605015, 61575028); National Science Foundation for Outstanding Youth Scholars of China (61622102); Fundamental Research Funds for the Central Universities (2016RCGD21, 2016RC26); Open Funds of IPOC (IPOC2016ZT03) and UWC (KFKT-2015102).

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Anti-dispersion phase-tunable microwave mixer based on a dual-drive dual-parallel Mach-Zehnder modulator

Abstract

1. Introduction

2. Theory and system structure

3. Experiment results

4. Conclusion

Funding

References and links

Cited By

Figures (7)

Equations (12)

Optics Express