Triple band frequency generator based on an optoelectronic oscillator with low phase noise

Zenghui Chen; Jian Dai; Yue Zhou; Feifei Yin; Tian Zhang; Jianqiang Li; Yitang Dai; Kun Xu

doi:10.1364/OE.25.020749

1. Introduction

Recently, multi-frequency band signal generator with high frequency and low phase noise is highly desirable in numerous electrical applications, such as the wireless communications, global positioning system and modern instrumentation and so on [1, 2]. Especially, the multi-function of weapon control, navigation and target detection in radar is realized simultaneously with demand of the phase-coherent multi-band frequency signals, which provides a novel strategy for other multi-band frequency electronic systems [3, 4]. However, inchoative multi-frequency is always obtained by multi-output frequency synthesizers consisted of a plenty of phase-locked loops (PLLs) and voltage-controlled oscillators (VCOs) [5, 6]. A large number of devices induced by PLLs and VCOs would result in complex systems and high costs. At present, the high frequency part of the multi-frequency is always in demand in the multi-function systems. Traditionally, it is generated by frequency multiplying through several stages of frequency doubling, which would bring about the serious deterioration of phase noise and the limited operated bandwidth [7–10]. In order to solve the problems of high cost, complex systems and poor phase noise, optoelectronic oscillator (OEO) with high frequency and low phase noise is being developed and receiving increasing attention [11–13].

Optoelectronic oscillator is a feedback loop where the continuous light energy can be converted into stable microwave or millimeter wave signals [14, 15]. It has many advantages compared with the conventional electrical schemes in the area of high frequency. For example, the high frequency signals with high spectral purity and low phase noise can be generated. Besides, an OEO with large tunable range can be achieved by employing a tunable microwave photonic filter, the stimulated Brillouin scattering effect or a tunable laser diode [4, 12, 16]. However, only one instantaneous frequency can be generated by the typical tunable OEO. By incorporating several independent OEOs and the tunable electrical band-pass filters, multi-frequency outputs can be realized easily as we anticipated, while the high cost and complex systems are not desirable in the age of the miniaturization and multifunction. Moreover, multi-frequency local signals also cannot be obtained simultaneously, which has limited the practical applications in wideband electronical systems [17, 18]. At present, a dual frequency can be achieved at the same time by using the OEO based on the phase-shifted fiber Bragg grating and ultra-narrow band optical notch filters. Nevertheless, the number of the output frequencies is limited to some extent while the devices with superior quality are required in the system [19]. Therefore, a generator based on the OEO that can simultaneously generate phase-coherent multi-frequency signals with low phase noise is extremely desired.

In this work, we proposed and experimentally demonstrated a triple band frequency generator based on an OEO. In the OEO, by biasing the intensity modulator at the MITP, the optical carrier is suppressed deeply. Then the modulated light is sent into the input port of the second modulator outside the OEO loop, which is biased near the MITP with a small deviation. As a result, the triple band frequency in C, Ku and K bands is realized simultaneously based on the proposed OEO system [8]. The three frequencies based on the rule of multiplication frequency are dependent in the characteristic of phase-coherent, which is very vital in multi-function integration systems. A proof-of-concept experiment has been carried out to verify the feasibility of the proposed scheme. The signals at 5.36, 16.08 and 21.44 GHz in C, Ku and K bands can be obtained after the second insufficient carrier suppression. The measured phase noise performance of the generated triple frequency signals is also investigated.

2. Principle of operation

The schematic of the proposed triple band frequency generator based on an optoelectronic oscillator is shown in Fig. 1. The optical signal from a tunable diode laser source is fiber coupled to the intensity modulator (MZM1) in the optoelectronic oscillator loop via a polarization controller (PC1), which ensures an efficient polarization direction before the input port of the MZM1. Then the modulated optical signal is divided into two parts by an optical coupler. One part is used as the optical source of the triple band frequency generating link outside the OEO loop. The other is sent to a single mode fiber (SMF), aiming to improve the phase noise performance of the optoelectronic oscillation loop. Owing to the optical to electrical conversion of the PD, the modulated signal is converted into electrical signal. The RF signal is further amplified by three electrical amplifiers and filtered by an electrical narrow-band band-pass filter (BPF), which helps to compensate the loss of the oscillation loop and select the oscillation frequency, respectively. Therefore, an efficient OEO is successfully constructed, and the optical source with only two ± 1st-order sidebands can be obtained by biasing the MZM1 at the minimum transmission point (MITP).

Fig. 1 Schematic of the proposed triple band frequency generation based on the OEO. LD: laser diode; PC: polarization controller; MZM: Mach-Zehnder modulator; OC: optical coupler; SMF: single-mode fiber; PD: photo detector; EA: electrical amplifier; BPF: bandpass filter; EC: electrical coupler; EDFA: erbium-doped fiber amplifier; PS: phase shifter; ESA: electrical spectrum analyzer; OSA: optical spectrum analyzer.

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The optical source and RF driving signal of the second intensity modulator (MZM2) are both provided by the OEO. A phase shifter incorporated before MZM2 is to adjust the phase deviation between two RF signals. However, the MZM2 is biased at one point near MITP, which has a small deviation compared with MITP, and the ± 1st-order and ± 2nd-order sidebands are generated. By beating at the PD, the fundamental microwave, the third frequency harmonic and frequency quadrupling signals are generated simultaneously in this system while the second frequency harmonic is suppressed deeply. The modulated signal outside the OEO loop is split two parts for observation via an optical couple (OC2). One part is monitored by an optical spectrum analyzer, and the other is converted into RF signal by a photo detector (PD2). Finally, the resulted fundamental signal, the third frequency harmonic and frequency-quadrupling are observed by an electrical spectrum analyzer. The erbium doped fiber amplifier (EDFA) here is applied to amplify the light before PD2.

In brief, the triple band frequency is realized by biasing the MZM1 and MZM2 at the MITP and near the MITP, respectively. The optical field of input light can be expressed mathematically as

E_{i n} (t) = E_{0} \cos (ω_{0} t)

where

E_{0}

is an optical field amplitude, and

ω_{0}

is the angular frequency of the laser diode.

The voltage of the RF driving signal for the first modulator can be expressed as

V_{1} (t) = V_{e 1} \cos (ω_{e} t + ϕ_{1})

where

V_{e 1}

,

ϕ_{1}

and

ω_{e}

are the amplitude, phase and angular frequency of the RF driving signal, respectively.

If the first modulator is operated at minimum transmission point ( $V_{b i a s 1} = V_{π 1}$ ), all even-order optical sidebands would be suppressed theoretically, the optical field at the output can be expressed approximately as

\begin{matrix} E_{o u t 1} (t) = E_{0} \cos (ω_{0} t) \cos [β_{1} \cos (ω_{e} t + ϕ_{1}) + \frac{π}{2}] \\ \approx - \frac{E_{0} J_{1} (β_{1})}{2} {\cos [(ω_{0} + ω_{e}) t + ϕ_{1}] + \cos [(ω_{0} - ω_{e}) t - ϕ_{1}]} \end{matrix}

where

β_{1} = π V_{e 1} / (2 V_{π 1})

is the phase modulation index and

V_{π 1}

is the half-wave voltage of the first modulator.

Similarly, the voltage of the microwave signal to drive the second modulator can be expressed as

V_{2} (t) = V_{e 2} \cos (ω_{e} t + ϕ_{2})

where

V_{e 2}

,

ω_{e}

and

ϕ_{2}

are the amplitude, angular frequency and phase of the RF driving signal, respectively. However, different from the first modulator, the second modulator is biased near the minimum transmission point with a small deviation (

V_{b i a s 2} = V_{π 2} + V_{π 2} \cdot 2 δ / π

). The ± 1st-order and ± 2nd-order sidebands can be generated, and the optical field at the output of the second modulator can be expressed as

\begin{matrix} E_{o u t 2} (t) = E_{o u t 1} (t) \cdot \cos [β_{2} \cos (ω_{e} t + ϕ_{2}) + \frac{π}{2} + δ] \\ \approx \frac{E_{0} J_{1} (β_{1}) J_{0} (β_{2}) \sin δ}{2} {\begin{array}{l} \cos [(ω_{0} + ω_{e}) t + ϕ_{1}] \\ + \cos [(ω_{0} - ω_{e}) t - ϕ_{1}] \end{array}} \\ + \frac{E_{0} J_{1} (β_{1}) J_{1} (β_{2})}{4} {\begin{array}{l} \cos [(ω_{0} + 2 ω_{e}) t + ϕ_{1} + ϕ_{2} + δ] \\ + \cos [(ω_{0} - 2 ω_{e}) t - ϕ_{1} - ϕ_{2} - δ] \\ + \cos (ω_{0} t) \cos (ϕ_{1} - ϕ_{2} - δ) \end{array}} \end{matrix}

where

β_{2} = π V_{e 2} / (2 V_{π 2})

is the phase modulation index of the second modulator. If the phase deviation

ϕ_{1} - ϕ_{2} - δ

introduced by the electrical phase shifter is equal to

π /2

, the

ω_{0}

component of the signal out of the second modulator will vanish, and the optical field at the output of the second modulator can be expressed as

\begin{matrix} E_{o u t 2} (t) \approx \frac{E_{0} J_{1} (β_{1}) J_{0} (β_{2}) \sin δ}{2} {\begin{array}{l} \cos [(ω_{0} + ω_{e}) t + ϕ_{1}] \\ + \cos [(ω_{0} - ω_{e}) t - ϕ_{1}] \end{array}} \\ + \frac{E_{0} J_{1} (β_{1}) J_{1} (β_{2})}{4} {\begin{array}{l} \cos [(ω_{0} + 2 ω_{e}) t + ϕ_{1} + ϕ_{2} + δ] \\ + \cos [(ω_{0} - 2 ω_{e}) t - ϕ_{1} - ϕ_{2} - δ] \end{array}} . \end{matrix}

Afterwards, the modulated optical signal is injected into a PD, and the final radio frequency signal is obtained and can be written as

V_{o u t} (t) = \frac{E_{0}^{2} J_{1}^{2} (β_{1})}{2} \cdot {\begin{array}{l} J_{0} (β_{2}) J_{1} (β_{2}) \sin δ \cos (ω_{e} t + ϕ_{2} + δ) \\ + J_{0}^{2} (β_{2}) \sin^{2} δ \cos (2 ω_{e} t + ϕ_{1}) \\ + J_{0} (β_{2}) J_{1} (β_{2}) \sin δ \cos (3 ω_{e} t + 2 ϕ_{1} + ϕ_{2} + δ) \\ + \frac{J_{1}^{2} (β_{2})}{4} \cos (4 ω_{e} t + 2 ϕ_{1} + 2 ϕ_{2} + δ) \end{array}} .

The amplitude of the resulted four frequency components from the output of PD2 are represented by $A_{1}$ , $A_{2}$ , $A_{3}$ and $A_{4}$ , respectively. They can be used for simplifying the model and theoretical analysis as shown as

\begin{array}{l} A_{1} = \frac{1}{2} E_{0}^{2} J_{1}^{2} (β_{1}) J_{0} (β_{2}) J_{1} (β_{2}) \sin δ = R \cdot J_{0} (β_{2}) J_{1} (β_{2}) \sin δ; \\ A_{2} = \frac{1}{2} E_{0}^{2} J_{1}^{2} (β_{1}) J_{0}^{2} (β_{2}) \sin^{2} δ = R \cdot J_{0}^{2} (β_{2}) \sin^{2} δ; \\ A_{3} = \frac{1}{2} E_{0}^{2} J_{1}^{2} (β_{1}) J_{0} (β_{2}) J_{1} (β_{2}) \sin δ = R \cdot J_{0} (β_{2}) J_{1} (β_{2}) \sin δ; \\ A_{4} = \frac{1}{8} E_{0}^{2} J_{1}^{2} (β_{1}) J_{1}^{2} (β_{2}) = R \cdot J_{1}^{2} (β_{2}) / 4 \end{array}

where

R = E_{0}^{2} J_{1}^{2} (β_{1}) / 2

is a parameter at a determined value of

β_{1}

(3.04). The amplitude

A_{4}

of the frequency-quadrupled signal is dependent on

β_{2}

, and it can achieve maximum value when

β_{2}

is fixed at a value (1.71). Therefore, both

J_{0} (β_{2})

and

J_{1} (β_{2})

can be worked out. In this case, the amplitude

A_{1}

,

A_{2}

and

A_{3}

are proportional to

\sin δ

,

\sin^{2} δ

and

\sin δ

, respectively. By biasing the MZM2 near the MITP with a tiny deviation, the second frequency harmonic can be suppressed deeply, while the fundamental microwave, the third frequency harmonic and frequency quadrupling signals are generated simultaneously in this system.

3. Experimental results and discussion

A proof-of-concept experiment for generation triple band frequency signals based on the configuration as shown in Fig. 1 has been implemented. A continuous wave laser diode (New Focus TLB 6700) with the power of 12 dBm is fixed at 1550 nm. The optical signal is modulated firstly by the 20 GHz intensity modulator (EOspace). A 3.2 km single mode fiber is inserted in the oscillation loop. Moreover, the PDs (U²T XPDV2120R) inside and outside the OEO loop have bandwidth of 40 GHz, and responsibility of 0.65 A/W at 1550 nm. In order to operate the oscillator, three RF amplifiers with total 70 dB gain are employed. The fundamental oscillation frequency is determined by the BPF, which has a 3-dB bandwidth of 20 MHz centered at 5.36 GHz. An optical spectrum analyzer (Yokogawa AQ6370C) with a resolution of 0.02 nm is employed to monitor the optical signal at the output of MZM2. In addition, the electrical signal after PD2 is injected into an electrical signal analyzer (Agilent N9030A) to measure its spectrum and phase noise.

To verify the feasibility of the proposed scheme, the MZM1 in the OEO loop is biased at the MITP. The fundamental microwave signal determined by the center frequency of BPF can be generated, and it can be measured by an electrical spectrum analyzer. It is clear in Fig. 2(a) that the oscillation frequency is 5.36 GHz, and no harmonic is observed in the figure. A zoom-in view of the spectrum with a span of 50 kHz is exhibited in Fig. 2(b). Simultaneously, the optical spectrum of the fundamental microwave signal is also monitored by an optical spectrum analyzer as shown in Fig. 3(a). By biasing MZM1 at the MITP, the optical carrier is suppressed deeply. As can be seen, the ± 1st-order sidebands are about 17.66 dB higher than the optical carrier. The weak ± 2nd-order and ± 3rd-order sidebands relative to the strong ± 1st-order sidebands can be ignored in experiment.

Fig. 2 Fundamental microwave signal generated by the OEO. (a) Electrical spectrum of the generated signal at 5.36 GHz. (b) Zoom-in view of the spectrum.

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Fig. 3 Optical spectra for generating triple band frequency. (a) Optical spectrum at the output of OC1. (b) Optical spectrum at the output of the MZM2. (c) Optical spectrum at the output of EDFA.

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To obtain the phase-coherent triple band frequency further, the modulated optical signal from the output of the MZM1 is sent into the second modulator. Being different from the first modulator, the modulator outside the OEO loop is biased near the MITP with a small deviation. A phase shifter is also used to insure the 90° phase difference between the two driven signals for the MZMs. The optical spectrum obtained at the output of the MZM2 is shown in Fig. 3(b). It can be seen that the two second-order optical sidebands are 30.88 dB larger than optical carrier. The ± 1st-order sidebands, which are the dominant component at the output of MZM1 have been suppressed to a large degree after passing through the MZM2. Besides, the optical output power is reduced for 12 dBm (0 dBm to −12 dBm) after passing through MZM2, which can be ascribed for two reasons. One is the existence of the insertion of the intensity modulator, the other is that the optical modulator biased near MITP resulted in the decrease of the output optical power. Then the optical signal is amplified by an EDFA, the corresponding optical spectrum was shown in Fig. 3(c).

Subsequently, the optical signal at the output of the second modulator is converted back to electrical domain by the photo detector. The electrical spectrum of the generated RF signal is shown in Fig. 4. As we can see, the phase-coherent signals at the 5.36, 16.08 and 21.44 GHz are obtained in the electrical spectrum. Though the power of three signals has small deviation to some extent, the triple band frequency in C, Ku and K band is achieved by adjusting the biasing point of two intensity modulator. Especially, by biasing the second modulator outside the OEO loop near the MITP with a small deviation ( $δ$ = 0.072), the second harmonic frequency signal is weakest when compared with other three signals. Therefore, the triple band frequency can be generated as we predicted in the proof-of-concept experiment. The three frequencies based on the rule of multiplication frequency are dependent in the characteristic of phase-coherent, which is very vital in multi-function integration systems. Moreover, the triple frequencies are dependent in frequency value, which can be stabilized simultaneously by stabilizing the fundamental frequency. Meanwhile, the generated frequencies can be coarsely tuned by using a tunable band-pass filter in the OEO loop and be flexibly tuned by combining these techniques including direct digital synthesis (DDS), PLL and mixing. Figures 5(a)-5(c) provided a zoom-in view of the spectral components at 5.36, 16.08 and 21.44 GHz, respectively.

Fig. 4 Electrical spectrum of the generated triple band frequency signals at 5.36, 16.08 and 21.44 GHz, respectively.

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Fig. 5 Zoom-in view of the electrical spectra (a-c) and phase noise measurements (d-f) of the generated 5.36 GHz, 16.08 GHz and 21.44 GHz signals, respectively.

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The phase noise performance of the triple band frequency signals is measured by an electrical spectrum analyzer with a phase noise measurement module. The phase noises of the generated signals at 5.36, 16.08 and 21.44 GHz at 10 kHz offset frequency are respectively −123.32 dBc/Hz, −113.68 dBc/Hz and −111.19 dBc/Hz, as shown in Figs. 5(d)-5(f). The triple frequency harmonic and frequency-quadrupling signals have a 9.64 dB and 12.13 dB phase noise degradation when compared with the fundamental signal, which is consistent with the theoretical prediction since the calculated degradation are given by $20 \log_{10}^{3} \approx 9.54$ dB and $20 \log_{10}^{4} \approx 12.0$ dB [20]. The low phase noise property of the OEO is delivered to the obtained third and fourth harmonic according to multiplexing rules. Besides, the phase noise of the phase-coherent triple frequencies in C, Ku and K bands can be further reduced by optimizing the noises induced by the optoelectronic oscillation loop. The spurious modes at integral multiples of 62.13 kHz can also be observed in Figs. 5(d)-5(f), which corresponds to the free spectral range of the proposed OEO or the inserted long fiber in the OEO loop (~3.2 km).

4. Conclusion

In conclusion, a novel approach for generating triple band frequency based on an OEO has been proposed and experimentally verified. The phase-coherent triple frequency in C, Ku and K bands is achieved by a full and an incomplete optical carrier suppression process inside and outside the OEO loop, respectively, which can be realized by biasing the intensity modulator inside the OEO at MITP and biasing another intensity modulator outside the OEO loop near the MITP with a small deviation. In the experiment, the phase-coherent triple frequency signals at 5.36, 16.08 and 21.44 GHz in C, Ku and K band are generated with the phase noises about −123.32, −113.68 and −111.19 dBc/Hz at 10 kHz offset frequency, respectively. Our proposed scheme shows a potential application in the fields of the multi-function and multi-band frequency electrical systems.

Funding

This work was supported in part by Natural National Science Foundation of China (NSFC) Program (61501051, 61625104 and 61431003).

Acknowledgments

Many thanks to Jinliang Liu for invaluable help in the laboratory testing of our scheme. Many thanks to Long Ye for useful discussions.

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Triple band frequency generator based on an optoelectronic oscillator with low phase noise

Abstract

1. Introduction

2. Principle of operation

3. Experimental results and discussion

4. Conclusion

Funding

Acknowledgments

References and links

Cited By

Figures (5)

Equations (8)

Optics Express